1. Field of the Invention
The present invention relates to a method of and a device for controlling a pulse width modulation (PWM) inverter for controlling its output voltages.
2. Description of the Prior Art
When driving an AC motor using a PWM inverter, high-frequency components included in each of the output voltages from the PWM inverter cause a magnetic attraction force and noise which is referred to as electromagnetic noise. Especially, since high-frequency components having the same frequency are continuously applied to the AC motor when the carrier frequency is constant, high electromagnetic noise is generated. In addition, AC motors have a plurality of natural frequencies that depend on their structure. Accordingly, if the output voltages from the PWM inverter include a high-frequency component with a frequency equal to any one of the plurality of natural frequencies of the AC motor which is controlled by the PWM inverter, vibrations generated in the AC motor are increased and the peak level of electromagnetic noise is increased.
To solve the above problem, Japanese Patent Application Laying Open (KOKAI) No. 7-177753 discloses a prior art control apparatus for controlling a PWM inverter. Referring now to FIG. 10, there is illustrated a schematic circuit diagram of the prior art control apparatus. In the figure, reference numeral 1 denotes a three-phase inverter circuit, 2 denotes a reference voltage vector generating unit for generating a reference voltage vector, 3 denotes a carrier signal generating unit for generating a carrier signal, 4 denotes a microcomputer, and 5 denotes a switching signal generating unit for generating a switching signal.
The three-phase inverter circuit 1 is comprised of a plurality of semiconductor switches Su, Sv, Sw, Su', Sv', and Sw', and a plurality of output terminals 1u, 1v, and 1w. Each of the plurality of semiconductor switches is comprised of a self-extinguishing switching element, such as a transistor, and a pair of diodes (not shown) connected in antiparallel to the switching element. The reference voltage vector generating unit 2 is comprised of an A/D converter 21, a ROM 22 for storing a k/f pattern or V/f pattern, a V/F converter 23, and a counter 24. The carrier signal generating unit 3 is comprised of a crystal oscillator 31 and an up/down counter 32. The switching signal generating unit 5 is comprised of comparators 51 to 53 and NOT gates 54 to 56.
Next, a description will be made as to the principle of PWM control of voltage-vector-selection type which the prior art control apparatus employs. Each of the three AC output phase voltages Vu, Vu, and Vw corresponding three phases U, V, and W, which differ in phase and respectively appear at the output terminals 1u, 1v, and 1w, can have any of two possible values, 0 and E, where E represents the voltage of a DC power supply 11 included in the three-phase inverter circuit 1. The switching status of the inverter in the case of Vu=E, Vv=0, and Vw=0 can be designated by a vector (E00). Normalizing this vector with E yields a vector (100), which is referred to as a voltage vector.
Since each of the three AC voltages Vu, Vv, and Vw can have any of two possible values, 0 and E, as previously mentioned, the three-phase inverter 1 can furnish any one of eight (=2.times.2.times.2) different voltage vectors. FIG. 11 shows such the eight different voltage vectors. In FIG. 11, the vertexes of the regular hexagon shown represent the voltage vectors V1[=(001)], V3[=(011)], V2[=(010)], V6[=(110)], V4[=(100)], and V5[=(101)], respectively. Each of the remaining two voltage vectors V0[=(000)] and V7[=(111)] whose line voltages are zero is referred to as a zero-voltage vector.
Next, a description will be made as to a method of controlling the output voltages when a reference voltage vector V* is located within a regular triangle having three vertexes defined by the two voltage vectors v4[=(100)] and V6[=(100)], and the zero-voltage vector V0[=(000)] or V7[=(111)], as shown in FIG. 12.
Assume that the reference voltage vector V* has an amplitude or magnitude of k and rotates clockwise at a frequency of .omega.. In order to make a vector representing the averages of the instantaneous amplitudes of the AC output voltages for a predetermined time period T agree with the reference voltage vector V*, the length of the locus, shaped like a circular arc, at the end of the reference voltage vector V*, has to be the same as that of the locus of the sum of the two voltage vectors V4[=(100)] and V6[=(100)], and the zero-voltage vectors V0[=(000)] and V7[=(111)]. Thus, the following equation (1) is established. ##EQU1## where .theta.=.omega.t, t4 and t6 represent the time durations of the voltage vectors V4 and V6, respectively. For simplicity, it is assumed that the magnitudes of the voltage vectors V4 and V6, which are measured from the origin shown in FIG. 12, are 1/.sqroot.3.
In addition, since the sum of the time durations of those voltage vectors V4 and V6, and zero-voltage vectors V0 and V7 is equal to the predetermined time period T, the following equation (2) is obtained: EQU t4+t6+t0+t7=T (2)
where t0 and t7 represent the time durations of the zero-voltage vectors V0 and V7, respectively, using the equations (1) and (2), the time durations of those voltage vectors V4 and V6, and zero-voltage vectors V0 and V7 are obtained and given by the following equations (3): EQU t4=T.multidot.k.multidot.sin (.pi./3-.theta.) EQU t6=T.multidot.k.multidot.sin .theta. EQU t0+t7=T.multidot.{1-k.multidot.sin (.pi./3+.theta.)} (3)
Accordingly, when the voltage vectors V4, V6, V0, and V7 are sequentially furnished for the time durations given by the above equations (3), the averages of the instantaneous amplitudes of the AC output voltages for the predetermined time period T are equivalent to the three coordinates of the reference voltage vector V*, respectively. Sharing the sum (t0+t7) between the zero-voltage vectors V0 and V7 will be mentioned below.
The above description is directed to pulse width modulation in a case where the phase angle .theta. of the reference voltage vector V* lies in a phase angle range of 0 to .pi./3, as shown in FIG. 12. However, varying the selection of two voltage vectors, other than zero-voltage vectors, each time the phase angle of the reference voltage vector V* changes by .pi./3 can offer a similar control operation even if the phase angle of the reference voltage vector V* lies in the range of .pi./3 to 2.pi..
Next, a description will be made to a sequential procedure of selecting two voltage vectors and two zero-voltage vectors with reference to FIG. 13. Two voltage vectors and the two zero-voltage vectors are selected sequentially according to any one of arrows shown in FIG. 13. For example, when the phase angle .theta. of the reference voltage vector V* lies in the range of 0 to .pi./3, the selection of those voltage vectors is carried out in the order of V0, V4, V6, and V7 during a predetermined time period T. The voltage vector selection is then carried out in the order of V7, V6, V4, and V0 during the next predetermined time period T. Thus, those voltage vectors V0, V4, V6, and V7 are selected sequentially in the aforementioned orders while the phase angle of the reference voltage vector V* lies in the range of 0 to .pi./3.
When the phase angle .theta. of the reference voltage vector v* increases and then shifts to a range of .pi./3 to 2.pi./3, another two voltage vectors, V2 and V6, and the two zero-voltage vectors are sequentially selected in the order of V0, V2, V6, V7, V6, V2, and V0 during two predetermined time periods 2T. In the selection sequence of voltage vectors which is carried out in such and order after the phase angle .theta. of the reference voltage vector V* shifts to the range of .pi./3 to 2.pi./3, the replacement of the voltage vector V4 with another voltage vector V2 is simply performed while the selection sequence of the remaining voltage vector V6 and the two zero-voltage voltage vectors is not changed. In addition, since the time durations of the voltage vectors V4 and V2 are almost zero when the phase angle .theta. of the reference voltage vector V* is .pi./3 or nearly .pi./3, as can be seen from FIG. 13, the output voltages do not change suddenly, even though the reference voltage vector V* moves from a phase angle range of .pi./3 to another adjacent phase angle range of .pi./3.
Next, a description will be given of a method of sharing the sum (t0 +t7) of the time durations of the two zero-voltage vectors V0 and V7 between them. FIG. 14(a) illustrates an example of the waveforms of output voltages when the time sharing of the sum (t0 +t7) between the two zero-voltage vectors V0 and V7 is maintained throughout all predetermined time periods. On the contrary, FIG. 14(b) illustrates an example of the waveforms of output voltages when the time sharing of the sum (t0 +t7) between the two zero-voltage vectors V0 and V7 is varied at predetermined time periods T. As can be seen from the waveform of a line voltage drawn at the bottom of FIG. 14(b), the average of the line voltage for one time period T is not varied while the interval between the two adjacent line voltage pulses is varied. Varying the time sharing of the sum of the time durations of the zero-voltage vectors between them causes the frequencies of high-frequency components included in each of the AC output voltages, i.e., output line voltages, to vary with time. Sharing the sum.tau.0=t0+t7 of the time durations of the zero-voltage vectors V0 and V7 between them can be done according to the following equations: EQU t0=.tau.0.multidot.b EQU t7=.tau.0.multidot.(1-b) (4)
where b is a random number that lies in the range of 0 to 1, and b also represents the value of a time sharing signal for determining the ratio of the time duration of the zero-voltage vector V0 to the sum of the time durations of the zero-voltage vectors V0 and V7.
In operation, the reference voltage vector generating unit 2 accepts and converts an analog reference frequency signal f into an equivalent digital signal by means of the A/D converter 21. When the ROM 22 receives the digital signal from the A/D converter 21, it furnishes a digital signal with a modulation factor of k according to the voltage/frequency pattern or k/f pattern stored therein. The reference frequency signal f is also converted into a series of pulses by the V/F converter 23. The counter 24 counts the series of pulses from the V/F converter 23 and then furnishes a digital signal representing the phase angle .theta. of a reference voltage vector, which has been obtained by intregrating the input reference frequency signal f.
On the other hand, the carrier signal generating unit 3 counts pulses included in a clock signal having a high frequency recieved from the crystal oscillator 31, by means of the up/down counter 32, and then generates a carrier signal having a triangular waveform. Simultaneously, the up/down counter 32 of the carrier signal generating unit 3 generates a clock signal which is synchronous with the time at which the up/down counter 32 switches between its counting up operation and its counting down operation. In other words, the carrier signal generating unit 3 generates a clock signal having a frequency two times as large as that of the carrier signal having a triangular waveform.
The microcomputer 4 carries out arithmetic operations in synchronization with the clock signal from the carrier signal generating unit 3 and then furnishes three reference phase voltage signals Vu*, Vv*, and Vw* as follows: When the microcomputer 4 accepts the modulation factor k and phase angle .theta. of the reference voltage vector V*, it divides the phase angle .theta. of the reference voltage vector V* by .pi./3 and then determines in which triangular region having a .pi./3 phase angle range, as shown in FIG. 13, the reference voltage vector V* lies according to the quotient of the phase angle .theta. divided by .pi./3, first. In other words, assuming that the quotient of the phase angle .theta. divided by .pi./3 defines the value of a region determination signal, the region determination signal can have any of six possible values, e.g., any one of the integers 0 to 5, which depends on the phase angle .theta. of the reference voltage vector V*. The six possible values of the region determination signal correspond to respective triangular regions (a) to (f) as shown in FIG. 13.
The microcomputer 4 then computes the time durations tb and tc of the two selected voltage vectors and the sum ta of the time durations of the two zero-voltage vectors according to the following equations (5) which are similar to the equations (3): EQU ta=T.multidot.{1-k.multidot.sin (.pi./3+.theta.)} EQU tb=T.multidot.k.multidot.sin (.pi./3-.theta.) EQU tc=T.multidot.k.multidot.sin .theta. (5)
The value b of the time sharing signal is determined by the execution of a random number generation function or by reading out a number from a table stored in a memory. After that, the microcomputer 4 determines the values of the three reference phase voltage signals Vu*, Vv*, and Vw*, using the value of the region de-termination signal, the time durations of the voltage vectors ta, tb, and tc, and the value b of the time sharing signal according to a relationship or table of FIG. 15 used for determining the timing of generation of the output voltages during one predetermined time period T. The microcomputer 4 then furnishes the reference phase voltage signals Vu*, Vv*, and Vw* to the switching signal generating unit 5.
In FIG. 15, Sa, Sb, and Sc used for determining the timing of generation of the three AC output voltages during one predetermined time period T, correspond to the reference phase voltage signals Vu*, Vv*, and Vw*, respectively. If the reference voltage vector V* lies in the triangular region (a), i.e., the phase angle of the reference voltage vector V* lies in the phase angle range (a) of 0 to .pi./3, Sa, Sb, and Sc are given by the following equations by using the time durations ta, tb, and tc and the value of the time sharing signal: EQU Sa=b.times.ta EQU Sb=Sa+tb EQU Sc=Sb+tc
If the reference voltage vector V* lies in the triangular region (b), Sa, Sb, and Sc are given by the following equations: EQU Sb=b.times.ta EQU Sa=Sa+tc EQU Sc=Sa+tb
Finally, the microcomputer 4 furnishes the reference phase voltage signals Vu*, Vv*, and Vw* to the switching signal generating unit 5 according to Sa, Sb, and Sc.
Referring next to FIG. 16, there is illustrated a graph showing an example of the waveforms of the three reference phase voltage signals Vu*, Vv*, and Vw* computed by the microcomputer, for the modulation factor k=0.8 and the value of the time sharing signal b=0.5.
Using the comparators 51 to 53, the switching signal generating unit 5 compares the amplitudes of the three reference phase voltage signals from the microcomputer 4 and that of the triangular-wave carrier signal from the carrier signal generating unit 3 and then generates and furnishes switching signals to the semiconductor switches Su, Sv, and Sw within the three-phase inverter circuit 1.
When the amplitude of each of the reference phase voltage signals is greater than that of the triangular-wave carrier signal, each of the comparators of the switching signal generating unit 5 generates a switching signal for turning on each of the semiconductor switches Su, Sv, and Sw of the three-phase inverter circuit 1. Each of the NOT gates 54 to 56 inverts the switching signal from each of the comparators so as to turn on or off each of the semiconductor switches Su', Sv', and Sw', so that those switches switch in a reverse way to the other switches Su, Sv, and Sw.
Thus the three-phase inverter circuit 1 furnishes three AC output voltages whose averages of the instantaneous amplitudes for one complete cycle of the clock signal agree with the three coordinates of the reference voltage vector V*, respectively. In addition, since the time durations of the two zero-voltage vectors are varied with time, according the value b of the time sharing signal, the frequencies of high-frequency components included in each of the output voltages are varied at random.
It was expected that the prior art control apparatus for controlling a PWM inverter constructed as mentioned above would spread the spectra of high-frequency components included in each of the AC output voltages over a wide range of frequencies. Hence high-frequency components of constant frequencies would be prevented from being continuously applied to an AC motor that is controlled by the PWM inverter by making the frequencies of high-frequency components of each of the output voltages vary at random, thereby reducing the magnitude of electromagnetic noise.
However, a problem with the prior art control apparatus is that the dispersion of the high-frequency components included in each of the AC output voltages can produce a high-frequency component having the same frequency as any one of the plurality of natural frequencies of the AC motor and therefore cause electromagnetic noise unbearable to users. Especially, if the high-frequency components included in each of the AC output voltages are spread out over a range of frequencies equal to or less than 1 kHz, the AC motor can cause electromagnetic noise whose tone quality varies as if the bearings were worn out. The problem is thus that the tone quality of the electromagnetic noise is reduced and it is therefore difficult to distinguish the electromagnetic noise from noise representing anomalies in the bearings.